Combined frequency-time domain power adaptation for CDMA communication systems

ABSTRACT

Practical transmission power adaptation in multicarrier code division multiple access (MC-CDMA) communications is using either a frequency domain technique or a time domain technique or a combined frequency and time domain technique in response to channel variations. With frequency domain power adaptation, the transmission power is allocated over the N′ (1≦N′≦N) strongest subcarriers rather than over all possible N subcarriers, where the strongest subcarriers are understood to exhibit the highest channel gains. A substantially optimal N′ can be chosen so that the average bit error rate (BER) is minimized. In the time domain power adaptation technique, transmission power is adapted so that the desired signal strength at the receiver output is maintained at a fixed level. In the combined time and frequency domain adaptation technique, the transmission power is first allocated over the N′ (1≦N′≦N) strongest subcarriers rather than over all possible N subcarriers and then it is adapted so that the desired signal strength at the receiver output is maintained at a fixed level.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application claims the benefit of priority from U.S. ProvisionalPatent Application Ser. No. 60/551,889 filed on Mar. 10, 2004. The aboveprovisional patent application is incorporated by reference herein inits entirety.

GOVERNMENT RIGHTS

The United States government may have certain rights in this invention.A portion of the work described herein was supported in part by theNational Science Foundation under NSF Grant ANS-03338788.

BACKGROUND OF THE INVENTION

1. Field of the Invention

This invention relates to a method and apparatus for wirelesscommunications and, more particularly, to adaptive power control forCDMA communication systems.

2. Description of the Related Art

Code division multiple access (CDMA) is one of several techniques formultiplexing wireless users. In CDMA systems, users are multiplexed overthe same wireless channel by using distinct spreading codes rather thanby using orthogonal frequency bands as in frequency division multipleaccess (FDMA) systems, or by using orthogonal time bands as in timedivision multiple access (TDMA) systems. In CDMA systems, all the userscan transmit simultaneously using the entire available transmissionspectrum.

Power control is an important part of these systems. Such power controlcan have a substantial impact on the capacity and apparent quality ofservice of the CDMA system. The need for power control in these systemsarises at least from the need to mitigate intercell interference thatarises from frequency reuse. In CDMA systems, a further need for powercontrol arises for minimizing intracell interference. For currentcode-division multiple-access (CDMA) cellular systems (IS-95),transmission power is adapted so as to maintain the same receiveddesired power level from all mobiles. When a wireless transmitter isprovided with suitable channel state information (CSI), adaptation ofthe transmission power in response to channel variations can be apowerful and efficient fading mitigation technique in wirelesscommunication systems.

In recent years, there has been considerable interest in multicarrier(MC) modulation techniques for high bit rate applications in fadingchannels. MC-CDMA modulation, a combination of frequency domainspreading and multicarrier modulation, is employed to achieve frequencydiversity and multiple access operation. MC-CDMA systems generally aredivided into two types: the first type encodes the original datasequence for a user via a spreading sequence and then a differentcarrier with each chip, and the second type spreads serial-to-parallelconverted data sequences using a given spreading code and then modulatesa different carrier with each of the data sequences. Se, for example, S.Hara et al., “Overview of multicarrier CDMA,” IEEE CommunicationsMagazine, pp. 126-133 (December 1997). A conventional MC-CDMAtransmitter allocates the available transmission power uniformly overall subcarriers.

In MC-CDMA systems, a form of power control or adaptation proposedwherein the transmitter uses only those subcarriers for which channelgains are higher than the given threshold level and truncates or turnsoff the other subcarriers while applying maximal ratio combining (MRC)at the receiver. See Zhu et al., “Performance of MC-CDMA systems usingcontrolled MRC with power control in Raleigh fading channel,” Elec.Lett., Vol. 36, pp. 752-53 (April 2000). This approach, however, resultsin transmission outages when the channel gains of all subcarriers arebelow the threshold level, a condition that is unacceptable for mostcommunication traffic that is intolerant for any number of reasons todelay. A power allocation algorithm for MC-CDMA with a projection matrixbased receiver was proposed where it was described that the optimalpower allocation coefficients are the components of the eigenvectorwhich corresponds to maximum eigenvalue of modified projection matrixorthogonal to the interference signal space. See Zhu et al., “Powerallocation algorithm in MC-CDMA,” Proc. IEEE ICC, pp. 931-35 (May 2002).This adaptation method requires the knowledge of all users' spreadingcodes and channel responses, which increases the system complexityespecially for larger number of users, making the implementationinfeasible for realistically sized systems expected in practice.

Although various adaptive power control techniques have been proposedfor MC-CDMA systems, none have presented a practical solution that canbe employed in currently expected communication systems based userquality of service demands and the size of the user community.

SUMMARY OF THE INVENTION

Practical transmission power adaptation in multicarrier code-divisionmultiple-access (MC-CDMA) communications is achieved in accordance withthe principles of the present invention using either a frequency domaintechnique or a time domain technique or a combined frequency and timedomain technique in response to channel variations. With frequencydomain power adaptation, the transmission power is allocated over the N′(1≦N′≦N) strongest subcarriers rather than over all possible Nsubcarriers, where the strongest subcarriers are understood to exhibitthe highest channel gains. A substantially optimal N′ can be chosen sothat the average bit error rate (BER) is minimized.

In the time domain power adaptation technique, transmission power isadapted so that the desired signal strength at the receiver output ismaintained at a fixed level. In the combined time and frequency domainadaptation technique, the transmission power is first allocated over theN′ (1≦N′≦N) strongest subcarriers rather than over all possible Nsubcarriers and then it is adapted so that the desired signal strengthat the receiver output is maintained at a fixed level.

In view of an average transmission power constraint placed on suchsystems, the frequency domain and the time domain power adaptationtechniques outperform the prior art non-adaptive techniques in whichconstant uniform power is allocated over all the N subcarriers. Inaddition, the combined frequency and time domain power adaptationtechnique provides a significant performance gain over the frequency ortime domain power adaptation techniques alone.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the invention may be obtained byreading the following description of specific illustrative embodimentsof the invention in conjunction with the appended drawings in which:

FIG. 1 shows a simplified block diagram of a CDMA wireless communicationsystem incorporating power adaptation in accordance with the principlesof the present invention;

FIG. 2 shows an exemplary power spectrum of a transmitted signal usingthe multi-carrier communication technique;

FIG. 3 shows a simplified block diagram of a portion of the modulator inthe transmitter from the system in FIG. 1;

FIG. 4 shows a simplified block diagram of a portion of the demodulatorin the receiver from the system in FIG. 1;

FIG. 5 shows a plot of the average bit error rate (BER) versus N′ forfrequency domain power adaptation in the system of FIG. 1 in accordancewith the principles of the present invention;

FIG. 6 shows a plot of the optimal number of N′ versus the number ofusers K for frequency domain power adaptation in the system of FIG. 1 inaccordance with the principles of the present invention;

FIG. 7 shows a plot of the average BER versus E_(b)/N₀ for all types ofpower adaptation applied in the system of FIG. 1 in accordance with theprinciples of the present invention; and

FIG. 8 shows a plot of the average BER versus the number of users K forall types of power adaptation applied in the system of FIG. 1 inaccordance with the principles of the present invention.

It should be noted that the appended drawings illustrate only typicalembodiments of this invention and are therefore not to be construed aslimiting of its scope, for the invention may admit to other equallyeffective embodiments. Where possible, identical reference numerals havebeen inserted in the figures to denote identical elements.

DETAILED DESCRIPTION

In the description that follows, certain terms are used interchangeably.The term “user” may be interchanged with the terms “mobile”, “mobilestation”, and “mobile user”. The term “base station” is also usedinterchangeably with the term “base”. The terms “transmit power” and“transmission power” are also used interchangeably. These usages are notintended to be limiting in any way.

The description below is organized as follows. A system model isintroduced and together with a derivation of thesignal-to-interference-plus-noise ratio (SINR) of the received signalswith maximum ratio combining (MRC). Next, the power adaptationtechniques are described and resulting bit error rate (BER) performancesare analyzed. Finally, numerical results are presented with a discussionof the performance improvements provided by present transmit poweradaptation techniques.

In the description below, transmission power adaptations are presentedalong the frequency domain, the time domain, and a combination of boththe frequency and time domains in an exemplary system utilizingquasi-synchronous uplink MC-CDMA communications with an MRC receiver.This exemplary system is presented as a framework within which tounderstand the principles of the present invention, as opposed tolimiting the applicability of this invention.

In frequency domain power adaptation, the transmission power isallocated uniformly over N′ (1>N′≦N) strongest subcarriers, whichexhibit the N′ largest channel gains, among the N available users on thesystem. When these N′ subcarriers exhibit channel gains that arerelatively high, the desired signal strength at the receiver increasesfor a given total transmission power. But it will be shown that a smallnumber of selected power adapted subcarriers N′ also leads to areduction of effective spreading gain which, in turn, helps to mitigatemultiple-access interference. The effect of selection of the number ofpower adapted subcarriers N′ on the performance of the frequency domainpower adaptation technique is described in detail below. One importanteffect is that there exists an optimal number of subcarriers N′ thatcontributes to a minimization of the average BER.

In the time domain power adaptation approach, the transmission power ofeach user is dynamically adapted so that the desired signal strength atMRC receiver output is maintained at a fixed level. In the combined timedomain and frequency domain power adaptation technique, the transmissionpower is first allocated over the N′ (1≦N′≦N) strongest subcarriers,rather than over all possible N subcarriers, and then the transmit poweris adapted so that the desired signal strength at the receiver output ismaintained at a fixed level. In the description that follows, it will beshown that the combined frequency-time domain power adaptation has asignificant performance gain over the power adaptation in only frequencydomain or in the time and that all these transmit power adaptationtechniques provide significant performance gains over a system in whichpower adaptation is not employed.

An exemplary MC-CDMA communication system is shown in FIG. 1, where Ksimultaneously active mobile stations 110 (i.e., mobile stations 110-1through 110-K) communicate with a base station 120. Relevant details ofthe transmit power spectrum, the transmitter architecture, and thereceiver architecture employed in the system of FIG. 1 are depicted inFIGS. 2-4. The system in FIG. 1 shows only one cell in a multi-cellsystem for ease of understanding, although the results derived hereinare applicable to multi-cell systems. The implications of amultiple-cell system can be analyzed and accounted for by simplyaccounting for an out-of-cell interference components. In this exemplarysystem, channel variations due to fading are assumed to be slow relativeto the data bit duration, and are also assumed to be independent fordifferent users.

At the transmitter in the system of FIG. 1, each mobile user employsN_(c) subcarriers and binary phase-shift-keying (BPSK) modulation.First, a user data sequence d_(k) k=1, 2, . . . , K, which output bydata source 111 at a rate of 1/T_(s), is serial-to-parallel (S/P)converted in transmitter 112 into F parallel streams. Each S/P convertedstream, one of which is shown in FIG. 3, is spread by a spreading code pin element 301 and then mapped or copied onto a subset N of the N_(c)subcarriers by mixer 302, where N_(c)=F×N. The mapped signals areadjusted by amplifier 303 using the α coefficients and then combinedtogether by summing junction 304 for output to and transmission byantenna 113. The power spectrum of an exemplary transmitted signal isshown in FIG. 2. Such an arrangement of subcarriers enables each of theN MC-CDMA subcarriers channel to be assumed independent. The exemplaryspectrum shown in FIG. 2 also shows the repetition of the same data biton each of the N subcarriers from one of the F parallel data streams ofthe S/P converted data sequence. It should be noted that F alsodetermines the frequency separation between neighboring subcarriersmodulated by the m^(th) data bit.

Generation of a transmit signal following the serial-to-parallelconversion is shown in FIG. 3. The transmitted signal corresponding tothe m^(th) data bit d_(k,m) of the user k can be expressed as follows:

$\begin{matrix}{{S_{k,m}(t)} = {\sum\limits_{n = 0}^{N - 1}{\sqrt{2\alpha_{k,n}S_{T}}{d_{k,m}(t)}{p_{k,n}(t)}{\cos\lbrack {{2\pi\; f_{c}t} + {2\pi\;{{nFt}/T}}} \rbrack}}}} & (1)\end{matrix}$where p_(k,0), p_(k,1), . . . , p_(k,N-1) represent the random spreadingsequence for mobile user k, and d_(k,m) is the m^(th) binary data bitwhich is serial-to-parallel converted with bit duration of T=FT_(s).S_(T) is the average transmission power, and α_(k,n) is the transmitterpower gain for the n^(th) subcarrier of mobile user k. In order to meetthe fixed average transmission power constraint,

$E\lbrack {\sum\limits_{n = 0}^{N - 1}\alpha_{k,n}} \rbrack$should be 1 for all k∈{0,1, . . . , K−1}. It is assumed that sufficientchannel state information (CSI) is available at both transmitter andreceiver. Techniques for acquiring and estimating CSI are well known inthe art.

For the purpose of the following analysis, it is assumed, without anyloss in generality or applicability of the results, that the channel isfrequency-selective, and that each subcarrier carrying the m^(th) databit experiences independent Rayleigh fading. The assumption ofindependent fading by each subcarrier is appropriate for channels whereF/T>>B_(c), for B_(c) representing the coherence bandwidth. The receivedsignal r(t) corresponding to bit m at the base station 120 can then bewritten as:

$\begin{matrix}{{{r(t)} = {{\sum\limits_{k = 0}^{K - 1}{\sum\limits_{n = 0}^{N - 1}{\sqrt{2\alpha_{k,n}S_{T}G_{k,n}}{d_{k,m}( {t - \tau_{k}} )}{p_{k,n}( {t - \tau_{k}} )}{\cos\lbrack {{2\pi\; f_{c}t} + {2\pi\;{{nFt}/T}} + \theta_{k,n}} \rbrack}}}} + {n(t)}}},} & (2)\end{matrix}$where τ_(k) is the delay of user k, which is assumed to be independentand uniformly distributed over a bit interval; μ_(k,n) is the channelinduced phase of user k at the n^(th) subcarrier, which is assumed to beindependent and uniformly distributed over [0,2π]; n(t) represents thewhite Gaussian noise with zero mean and two-sided power spectral densityN₀/2; G_(k,n) is an exponentially distributed random variablerepresenting the channel power gain for user k at the n^(th) subcarrier,whose probability density function is given by:

$\begin{matrix}{{{P_{G_{k,n}}(g)} = {\frac{1}{\Omega_{0}}{\mathbb{e}}^{{- g}/\Omega_{0}}}},} & (3) \\{where} & \; \\{\Omega_{0} = {{E\lbrack G_{k,n} \rbrack}.}} & (4)\end{matrix}$It is assumed that Ω₀ is normalized to unity. Due to the complex natureof the transmit signal structure, the theoretical analysis herein isfocused only on the m^(th) data bit carried by N subcarriers out of m,m+1, . . . , m+F data bits carried by N_(c) subcarriers.

At the base station 120, the users' signals are received by the antenna.The received signal is supplied to each of the receivers 124corresponding to users k=1, 2, . . . , K. Receiver 124 is depicted inmore detail in FIG. 4 for extracting the communications from user 1 (thesubscript is one less than the user identifier k). In FIG. 4, thereceived signal r(t) is propagated down each of N substantially similarpaths. For simplicity, the operation along path n is described herein.

Mixer 401 extracts the signal on the n^(th) subcarrier. This signal isthen despread by correlator 402 using spreading code p_(0,n). Gain ofthe despread signal is then adjusted by amplifier 403. Summing junction404 combines the subcarrier constituent signals for the correspondingdata bit together. After integration over a bit period by element 405,the recovered data bit {circumflex over (d)}₀ is output by switchingelement 406.

It is assumed that substantially exact synchronization (otherwise knownin the art as perfect phase correction) with the desired user can beobtained. For the m^(th) bit from user 1, the decision variable is givenby:

$\begin{matrix}\begin{matrix}{{\hat{d}}_{0} = {\sqrt{\frac{2}{T}}{\int_{T}{{r(t)}{\sum\limits_{m = 0}^{N - 1}{\sqrt{\beta_{0,m}}{p_{0,m}( {t - \tau_{0}} )}{\cos\lbrack {{2\pi\; f_{c}t} +} }}}}}}} \\{ {{2\pi\;{{mFt}/T}} + \theta_{0,m}} \rbrack\mspace{11mu}{\mathbb{d}t}} \\{= {{\sum\limits_{n = 0}^{N - 1}{\sqrt{\alpha_{0,n}\beta_{0,n}\; G_{0,n}E_{b}}{d_{0}\lbrack m\rbrack}}} + I_{MAI} + \eta}}\end{matrix} & (5)\end{matrix}$where the subscript of the data bit is one less than the user identifierk (here k=1), β_(k,n) is the receiver power gain for the n^(th)subcarrier of user k, and bit energy E_(b)≡S_(T)T. The first term inEquation (5) is the desired signal term. The second term labeled I_(MAI)is the multiple-access interference term induced by the other K−1 activeusers, and the third term η is the white Gaussian noise. I_(MAI) and ηare independent random variables with mean zero and variances

$\begin{matrix}{{E\lbrack I_{MAI}^{2} \rbrack} = {\frac{E_{b}}{2}{\sum\limits_{k = 1}^{K - 1}{\sum\limits_{n = 0}^{N - 1}{\beta_{0,n}{E\lbrack {\alpha_{k,n}G_{k,n}} \rbrack}}}}}} & (6) \\{{{E\lbrack \eta^{2} \rbrack} = {\frac{N_{0}}{2}{\sum\limits_{n = 0}^{N - 1}\beta_{0,n}}}},} & (7)\end{matrix}$respectively. Therefore, the SINR, Γ, for the desired user 1 can begiven by

$\begin{matrix}{\Gamma = {\frac{( {\sum\limits_{n = 0}^{N - 1}\sqrt{\alpha_{0,n}\beta_{0,n}\; G_{0,n}}} )^{2}}{{\frac{1}{2}{\sum\limits_{k = 1}^{K - 1}{\sum\limits_{n = 0}^{N - 1}{\beta_{0,n}{E\lbrack {\alpha_{k,n}G_{k,n}} \rbrack}}}}} + {\frac{N_{0}}{2E_{b}}{\sum\limits_{n = 0}^{N - 1}\beta_{0,n}}}}.}} & (8)\end{matrix}$

When channel state information (CSI) is not available at thetransmitter, transmission power is not adaptively allocated. Instead itis allocated uniformly over total N subcarriers. This is commonlyunderstood to be the case in prior art systems. Performance of such anon-adaptive transmission technique is presented below for the purposeof subsequent performance comparison with power adaptation techniquesdescribed below in accordance with the principles of the presentinvention. In this case, the transmitter power gain, α_(k,n), for user kis given by:α_(k,n) ^(no)=1/N, n=0,1, . . . , N−1.  (9)

It is reasonable to consider the maximal ratio combining (MRC) receiverin FIGS. 1 and 4 as a diversity combining method. The correspondingreceiver power gain, β_(k,n), for user k is:β_(k,n) ^(no) =G _(k,n) , n=0,1, . . . , N−1.  (10)By substituting Equation (9) and Equation (10) in Equation (8), oneobtains the SINR, Γ_(no), for the prior art non-adaptive transmissiontechnique as follows:

$\begin{matrix}{\Gamma_{no} = {v_{no}{\sum\limits_{n = 0}^{N - 1}G_{0,n}}}} & (11) \\{where} & \; \\{v_{no}\overset{\Delta}{=}{\lbrack {\frac{K - 1}{2} + {\frac{N}{2}\frac{N_{0}}{E_{b}}}} \rbrack^{- 1}.}} & (12)\end{matrix}$Using a Gaussian approximation based on the assumption that theinterference plus noise in Equation (5) is Gaussian with zero mean and avariance of E[I_(MAI) ²]+E[η²], it is possible to calculate the biterror rate (BER). Accordingly, the average BER for the non-adaptivetransmission technique is given by

$\begin{matrix}{{\overset{\_}{P}}_{b} = {\int_{0}^{\infty}{{Q( \sqrt{v_{no}g} )}P_{G_{0}{(g)}}{\mathbb{d}g}}}} & (13)\end{matrix}$where P_(b) is the probability of a bit error and parameters Q(x) andG_(k) are defined as follows:

$\begin{matrix}{{{Q(x)}\overset{\Delta}{=}{\frac{1}{\sqrt{2\pi}}{\int_{x}^{\infty}{{\mathbb{e}}^{{- t^{2}}/2}{\mathbb{d}t}}}}},{x \geq 0}} & (14) \\{and} & \; \\{{G_{k}\overset{\Delta}{=}{\sum\limits_{n = 0}^{N - 1}G_{k,n}}},} & (15)\end{matrix}$respectively. The probability density function of G_(k) is given by:

$\begin{matrix}{{{P_{G_{k}}(g)} = \frac{g^{N - 1}{\mathbb{e}}^{- g}}{( {N - 1} )!}},\mspace{14mu}{g \geq 0.}} & (16)\end{matrix}$Then by substituting Equation (16) in Equation (13) and using the resultof known techniques in Proakis, Digital Communications, 3^(rd) Ed., p.781, Eq. (14-4-15) (McGraw-Hill 1995), it is shown that the average BERin Equation (13) can be written as:

$\begin{matrix}{{\overset{\_}{P}}_{b} = {( \frac{1 - \mu}{2} )^{N}{\sum\limits_{m = 0}^{N - 1}{\begin{pmatrix}{N - 1 + m} \\m\end{pmatrix}( \frac{1 + \mu}{2} )^{m}\mspace{14mu}{where}}}}} & (17) \\{\mu\overset{\Delta}{=}{\sqrt{\frac{v_{no}}{2 + v_{no}}}.}} & (18)\end{matrix}$

Power adaptation in accordance with the principles of the presentinvention can be understood in the context of FIG. 1. In the basestation 120, the received signal r(t) is supplied to channel estimationelement 121 to obtain an estimate of the signal strength and, thereby,the channel power gains G_(k). The estimated channel gains are thenordered in ordering element 122. Ordering is performed for the estimatedchannel gains in decreasing order. With the channel gains ordered, it ispossible in element 123 to calculate the amplifier power gains α and βfor the transmitters and receivers in the system. Once calculated by thetime domain and frequency domain power adaptation techniques describedbelow in more detail, the amplifier power gains α and β are supplied tothe system transmitters and receivers. A feedback channel is shown forsupplying the transmitter amplifier power gains a from the calculationelement 123 to the K user mobile station transmitters.

While feedback channel is shown as a solid line in the figure, it isunderstood that this depiction was done for ease of understanding. Thefeedback channel can be realized as a real or logical or virtualwireless channel received by each of the mobile station antennas. Asintended to be realized, the transmitter amplifier power gains α arereceived by the related mobile station on its respective antenna and arethen processed and applied to the user's associated transmit poweramplifiers 303 in transmitter 112, wherein the transmit power of eachsubcarrier is adjusted to the level related to the transmitter amplifierpower gain α for that user.

As stated previously, the feedback channel can be a real channel such asa separate dedicated signaling or control channel allocated within thecommunication protocol. Alternatively, the feedback channel can berealized as a logical or virtual channel by using, for example, anoverhead portion or a payload portion of X symbols in a datatransmission sequence from the base station for transmitting the adaptedtransmit power gain levels. This information could be recovered by thetransmitter and applied to the corresponding transmit power amplifier.It is contemplated that the feedback signal would include the sequenceof adapted transmit power amplifier gain levels together with anindicator or index associating each level with its particular mobileuser station.

In this section, the transmission power is allocated in accordance withthe principles of the present invention uniformly over only the N′(1≦N′≦N) subcarriers that have highest channel gains rather than bydistributing it over all possible N subcarriers as done in thenon-adaptive transmission prior art technique described above. Thetransmitter power amplifier gains with such a frequency domain poweradaptation technique are calculated in element 123 as follows:

$\begin{matrix}{\alpha_{k,n}^{f} = \{ \begin{matrix}{{1/N^{\prime}},} & {{{if}\mspace{14mu} G_{k,n}} > G_{k,N^{\prime}}^{\prime}} \\{0,} & {otherwise}\end{matrix} } & (19)\end{matrix}$where G′_(k,0)≧G′_(k,1)≧ . . . ≧G′_(k,N-1) are the order statisticsobtained by arranging the estimated channel gain for the N subcarriersof user k,

$\begin{matrix}{\beta_{k,n}^{f} = \{ \begin{matrix}{G_{k,n},} & {{{if}\mspace{14mu} G_{k,n}} > G_{k,N^{\prime}}^{\prime}} \\{0,} & {{otherwise}.}\end{matrix} } & (20)\end{matrix}$in decreasing order. Similarly, the receiver power amplifier gainscalculated using the frequency domain power adaptation are given as:

{G_(k, n)}_(n = 1)^(N − 1),It should be understood that the receiver combining formation inEquation (20) corresponds to the signal being transmitted over only N′strongest subcarriers.

Employing the frequency domain power adaptation, one can show that:

$\begin{matrix}{{E\lbrack {\alpha_{k,n}^{f}G_{k,n}} \rbrack} = {\frac{1}{N}{( {1 + {\sum\limits_{l = {N^{\prime} + 1}}^{N}{1/l}}} ).}}} & (21)\end{matrix}$It follows from Equations (8), (19), (20), and (21) that thesignal-to-interference-plus-noise ratio (SINR), Γ_(f), for the frequencydomain power adaptation is given by:

$\begin{matrix}{\Gamma_{f} = {v_{f}Z_{0}\mspace{14mu}{where}}} & (22) \\{v_{f}\overset{\Delta}{=}{\lbrack {{\frac{( {K - 1} )N^{\prime}}{2N}( {1 + {\sum\limits_{l = {N^{\prime} + 1}}^{N}{1/l}}} )} + {\frac{N^{\prime}}{2}\frac{N_{0}}{E_{b}}}} \rbrack^{- 1}\mspace{14mu}{and}}} & (23) \\{Z_{0}\overset{\Delta}{=}{\sum\limits_{n = 0}^{N^{\prime} - 1}{G_{0,n}^{\prime}.}}} & (24)\end{matrix}$The probability density function of Z₀ is given by:

$\begin{matrix}{{P_{Z_{0}}(g)} = {{\begin{pmatrix}N \\N^{\prime}\end{pmatrix}\lbrack {\frac{g^{N^{\prime} - 1}e^{- g}}{( {N^{\prime} - 1} )!} + {\sum\limits_{l = 1}^{N - N^{\prime}}{( {- 1} )^{N^{\prime} + l - 1}\begin{pmatrix}N \\N^{\prime}\end{pmatrix}( \frac{N^{\prime}}{l} )^{N^{\prime} - 1} \times {e^{- g}( {e^{\frac{- l}{N^{\prime}}g} - {\sum\limits_{m = 0}^{N^{\prime} - 2}{\frac{1}{m!}( {{- \frac{l}{N^{\prime}}}g} )^{m}}}} )}}}} \rbrack}.}} & (25)\end{matrix}$Thus, the average BER with the frequency domain power adaptation isgiven by,

$\begin{matrix}{{\overset{\_}{P}}_{b} = {\int_{0}^{\infty}{{Q( \sqrt{v_{f}g} )}P_{Z_{0}{(g)}}{{\mathbb{d}g}.}}}} & (26)\end{matrix}$By substituting Equation (25) into Equation (26) and using a publishedresult, we get the following for average BER for the frequency domainpower adaptation:

$\begin{matrix}{{\overset{\_}{P}}_{b} = {\frac{\begin{pmatrix}N \\N^{\prime}\end{pmatrix}}{\pi}{\int_{0}^{\pi/2}{\frac{\sum\limits_{l = 0}^{\;{N - N^{\prime}}}{( {- 1} )^{l}\begin{pmatrix}{N - N^{\prime}} \\l\end{pmatrix}( {1 + \frac{1}{N^{\prime}} + \frac{v_{f}}{2\;\sin^{2}\phi}} )^{- 1}}}{( {1 + \frac{v_{f}}{2\;\sin^{2}\phi}} )^{N^{\prime} - 1}}{{\mathbb{d}\phi}.}}}}} & (27)\end{matrix}$

In this section, the transmission power is adapted in the time domainonly. According to this technique, the transmission power is distributedover all N subcarriers and, in contrast to the prior art non-adaptivetechniques, the transmission power is dynamically adapted for eachsymbol in the time domain so that the desired signal strength at amaximal ratio combining receiver maintains fixed desired level. Thetransmitter and the receiver power gains with such a power adaptationtechnique are random variables calculated in element 123 as follows:α_(k,n) ^(t)=α_(k) ^(t) /N, n=0,1, . . . , N−1  (28)andβ_(k,n) ^(t) =G _(k,n) , n=0,1, . . . , N−1,  (29)respectively. In order to satisfy the average transmission powerconstraint that applies to most commercial wireless systems, theexpected value of the transmitter amplifier gain levels E[α_(k) ^(t)]should be substantially 1.

In order to maintain the received power of the desired signal at a fixedlevel, G_(R) ^(t), the transmitter amplifier power gain is adjustedaccording to the following equation:α_(k) ^(t) =G _(R) ^(t) /G _(k)  (30)where the channel gain G_(k) is defined in Equation (15). It thenfollows from the average power constraint of the system and Equation(16) that G_(R) ^(t), is given by:

$\begin{matrix}{{G_{R}^{t} = {\frac{1}{E\lbrack {1/G_{k}} \rbrack} = {\lbrack {\frac{1}{( {N - 1} )!}{\int_{0}^{\infty}{g^{N - 2}{\mathbb{e}}^{- g}\ {\mathbb{d}g}}}} \rbrack^{- 1} = {N - 1}}}},} & (31)\end{matrix}$where, in the last step, integral tables can be used.

It can be shown that,

$\begin{matrix}{{E\lbrack {\alpha_{k,n}^{t}G_{k,n}} \rbrack} = {\frac{N - 1}{N^{2}}.}} & (32)\end{matrix}$Also from Equations (28), (30), and (31), the transmitter amplifier gainlevels can be expressed as,

$\begin{matrix}{{\alpha_{k,n}^{t} = \frac{N - 1}{{NG}_{k}}},{n = 0},1,\ldots\mspace{11mu},{N - 1.}} & (33)\end{matrix}$Substituting Equations (29), (32), and (33) into Equation (8) yields thesignal-to-interference-plus-noise ratio (SINR), Γ_(t), for the timedomain power adaptation technique as,

$\begin{matrix}{\Gamma_{t} = {\lbrack {\frac{K - 1}{2N} + {\frac{N}{2( {N - 1} )}\frac{N_{0}}{E_{b}}}} \rbrack^{- 1}.}} & (34)\end{matrix}$It should be noted that the SINR Γ_(t) does not fluctuate with thechannel fading, since the transmitter adapts its power levels tomaintain a constant SINR at the receiver. The average BER realized withthe time domain power adaptation is given by,P _(b) =Q(√{square root over (Γ_(t))})  (35)

The combined frequency-time domain power adaptation is a two-stepprocess in which frequency domain power adaptation is performed prior tothe time domain power adaptation. In the frequency domain poweradaptation, the transmission power is allocated over only the subset ofN′ strongest subcarriers, where 1≦N′≦N. Power adaptation is theperformed in the time domain to maintain the desired user signalstrength at a desired fixed level. With such a combined adaptationtechnique, the transmitter and the receiver power amplifier gains aregiven by:

$\begin{matrix}{\alpha_{k,n}^{ft} = \{ {\begin{matrix}{{\alpha_{k}^{ft}/N^{\prime}},} & {{{if}\mspace{14mu} G_{k,n}} > G_{k,N^{\prime}}^{\prime}} \\{0,} & {otherwise}\end{matrix},{and}} } & (36) \\{\beta_{k,n}^{ft} = \{ {\begin{matrix}{G_{k,n},} & {{{if}\mspace{14mu} G_{k,n}} > G_{k,N^{\prime}}^{\prime}} \\{0,} & {otherwise}\end{matrix},} } & (37)\end{matrix}$respectively.In order to maintain the received power of the desired mobile stationsignal at a desired fixed level, the transmitter power amplifier levelsare given as:α_(k) ^(ft) =G _(R) ^(ft) /Z _(k),  (38)where Z_(k) is defined in Equation (24). To meet the average powerconstraint in these systems, E[α_(k) ^(ft)] is preferably equal tounity. Hence,

$\begin{matrix}{{G_{R}^{ft} = {\frac{1}{E\lbrack {1/Z_{k}} \rbrack} = \Psi^{- 1}}}{where}} & (39) \\{\Psi\overset{\Delta}{=}{{E\lbrack {1/Z_{k}} \rbrack} = {\int_{0}^{\infty}{\frac{1}{g}{P_{Z_{k}}(g)}\ {{\mathbb{d}g}.}}}}} & (40)\end{matrix}$With Equations (36), (37), and (39), it is possible to show that:

$\begin{matrix}\begin{matrix}{{E\lbrack {\alpha_{k,n}^{ft}G_{k,n}} \rbrack} = {\sum\limits_{m = 0}^{N - 1}{{E\lbrack {{{\alpha_{k,n}^{ft}G_{k,n}}❘G_{k,n}} = G_{k,m}^{\prime}} \rbrack}{\Pr( {G_{k,n} = G_{k,m}^{\prime}} )}}}} \\{= {\frac{1}{N}{\sum\limits_{m = 0}^{N^{\prime} - 1}{E\lbrack {\alpha_{k}^{ft}{G_{k,m}^{\prime}/N^{\prime}}} \rbrack}}}} \\{= {\frac{G_{R}^{ft}}{{NN}^{\prime}}{\sum\limits_{m = 0}^{N^{\prime} - 1}{E\lbrack {G_{k,m}^{\prime}/Z_{k}} \rbrack}}}} \\{= {\frac{1}{{NN}^{\prime}\Psi}.}}\end{matrix} & (41)\end{matrix}$By substituting Equations (36), (37), and (41) into Equation (8), it ispossible to show that the SINR, Γ_(ft), for the frequency-time domainpower adaptation technique as:

$\begin{matrix}{\Gamma_{ft} = {\lbrack {\frac{K - 1}{2N} + {\frac{N^{\prime}\Psi}{2}\frac{N_{0}}{E_{b}}}} \rbrack^{- 1}.}} & (42)\end{matrix}$Therefore, the average BER is given by:P _(b) =Q(√{square root over (Γ_(ft))})  (43)

FIG. 5 shows a plot of the average bit error rate (BER) versus N′ forfrequency domain power adaptation in the system of FIG. 1 in accordancewith the principles of the present invention. The number of subcarriersN was selected to be 8 and the bit energy to noise spectral densityratio, E_(b)/N₀, was chosen to be 10 dB. The number of users, K, isvaried from 2 in curve 502 to 5 in curve 501. In FIG. 5, the average BERin Equation (26) is depicted as a function of N′, from which it can beobserved that there exists a value of N′ that minimizes the average BER.This set of graphs indicates that, for given system parameters, theaverage BER with the frequency domain power adaptation can be minimizedby appropriately choosing the number of N′.

FIG. 6 shows a plot of the optimal number of N′ versus the number ofusers K for frequency domain power adaptation in the system of FIG. 1 inaccordance with the principles of the present invention. The number ofsubcarriers N is again selected to be 8. In FIG. 6, the optimal N′ isplotted as a function of the number of active users K for several valuesof E_(b)/N₀. Curve 604 utilizes E_(b)/N₀ of 5 dB; curve 603 utilizesE_(b)/N₀ of 10 dB; curve 602 utilizes E_(b)/N₀ of 15 dB; and curve 601utilizes E_(b)/N₀ of 20 dB. The plot shows that the optimal value of N′increases with an increasing number of users K. This is because a higherspreading gain is required to mitigate the multiple access interferencefor larger numbers of users, K (i.e., interference limited region). Onthe other hand, a smaller number of optimal subcarriers N′ yields betterperformance for a smaller number of users K (i.e., noise-limited region)because, in order to mitigate the channel fading impairment, diversitygain obtained by reducing N′ is needed more than spreading gain. Itshould be noted that N′=1, which corresponds to selection diversity atthe mobile transmitter, is optimal for single-user case, where K=1.

FIG. 7 shows a plot of the average BER versus E_(b)/N₀ for all types ofpower adaptation applied in the system of FIG. 1 in accordance with theprinciples of the present invention. The number of available subcarriersN is selected to be 8 and the number of users K is selected to be 4. Theaverage BERs for several adaptation techniques are compared in FIG. 7,wherein the optimal N′ that minimizes the average BER was used for boththe frequency domain and the frequency-time domain power adaptations. Itshows that the combined frequency-time domain power adaptation (curve704) has a significant performance gain over the non-adaptive technique(curve 701). While the time domain power adaptation (curve 703) and thefrequency domain power adaptation (curve 702) show better performancethan the non-adaptive technique, they show complementary performance indifferent Eb/N₀ regions. For lower Eb/N₀ in the so-called noise-limitedregion, the frequency domain power adaptation outperforms thetime-domain power adaptation. For higher Eb/N₀ in the so-calledinterference-limited region, the time domain power adaptationoutperforms the frequency domain power adaptation. Therefore, FIG. 7indicates that for MC-CDMA communications, joint adaptation of thetransmission power in the frequency-time domain makes more efficient useof the available transmission power than power adaptation in thefrequency domain or the time domain alone.

FIG. 8 shows a plot of the average BER versus the number of users K forall types of power adaptation applied in the system of FIG. 1 inaccordance with the principles of the present invention. The number ofsubcarriers N was selected to be 8 and the signal to noise ratioE_(b)/N₀ was chosen to be 15 dB. FIG. 8 shows the average BER versus Kwith several adaptation techniques. Again the combined frequency-timedomain power adaptation in curve 801 yields much better performance thanthe non-adaptive transmission technique in curve 804 for all numbers ofusers, K. The performance gain becomes more pronounced as the number ofusers K decreases. The time domain power adaptation technique in curve802 outperformed the frequency domain power adaptation technique incurve 803 under these conditions. Such performance gain can betranslated into a reduction of the total number of subcarriers requiredto achieve a target BER for a given number of users, which in turn meansa reduction of required system bandwidth.

The power adaptation techniques described above are based on knowledgeof the channel state information on each of the subcarriers. Inexperimental practice, the performance of the power adaptationtechniques can be negatively affected if the channel state estimation isnot reliable. Therefore, the transmission power adaptations in MC-CDMAcommunication systems which require the transmitter to have reliable apriori CSI about the subcarriers can be applied to systems withslowly-varying channel characteristics, such as cellular systems forpedestrian or nomadic environments, wireless local area networks (WLANor WiFi), or wireless local loop (WLL) systems.

The description above has set forth details about a novel technique foradapting transmission power in MC-CDMA communication systems in thefrequency domain, in the time domain, and in a combination of the timeand frequency domains. For frequency domain power adaptation, thetransmission power has been allocated to only the N′ strongestsubcarriers. In time domain power adaptation, the transmission power isadjusted to maintain the signal strength at a fixed level. A combinationof the frequency domain power adaptation and the time domain poweradaptation outperforms the non-adaptive technique for noise-limited andinterference-limited regions, respectively. The combined adaptation ofthe transmission power in the combined frequency-time domain was shownto significantly outperform the power adaptation in only frequency ortime domain as well as over the non-adaptive technique.

While the foregoing is directed to embodiments of the present invention,other and further embodiments of the invention may be devised withoutdeparting from the basic scope thereof, and the scope thereof isdetermined by the claims that follow.

1. Method for power adaptation in a multicarrier code division multipleaccess communication system in which a plurality of mobile stationscommunicate with at least one base station, each mobile stationincluding an adaptive transmitter that includes a plurality ofcontrollable gain amplifiers, the at least one base station including aplurality of adaptive receivers, each receiver corresponding to one ofthe plurality of mobile stations and including a plurality ofcontrollable gain amplifiers, the method comprising the steps of:receiving signals from the mobile stations; estimating channel powergain for each mobile station in response to the received mobile stationsignals, where the channel power gain G_(k,n) is for an n^(th)subcarrier of mobile station k, for k=1, 2, . . . , K and n=1, 2, . . ., N; ordering received signals from the mobile stations on a decreasingorder basis for the corresponding estimated channel power gain from themobile station associated with a highest channel gain to the mobilestation associated with a lowest channel gain; determining both aplurality of power amplifier gain levels α_(k,n) ^(t) for thecontrollable gain amplifiers in the adaptive transmitter associated witheach mobile station and a plurality of power amplifier gain levelsβ_(k,n) ^(t) for the controllable gain amplifiers in the adaptivereceiver associated with each mobile station in response to the orderedchannel power gains, wherein the adaptive transmitter power amplifiergain levels and the adaptive receiver power amplifier gain levels aredynamically adapted to the estimated channel power gains for each symbolfrom the mobile stations in the time domain for distributing availabletransmission power over all N subcarriers to maintain a received signalstrength for signals received at the base station at a predeterminedlevel.
 2. The method as defined in claim 1 wherein the determining stepfurther includes the steps of: calculating the adaptive transmitterpower amplifier gain levels as follows,α_(k,n) ^(t)=α_(k) ^(t) /N, n=1,2, . . . , N, where${\alpha_{k}^{t} = {G_{R}^{t}/G_{k}}},{G_{k} = {\sum\limits_{n = 0}^{N - 1}\; G_{k,n}}}$and G_(R) ^(t) is the predetermined level; and calculating the adaptivereceiver power amplifier gain levels as follows:β_(k,n) ^(t)=G_(k,n), n=1,2, . . . , N.
 3. The method as defined inclaim 2 wherein the predetermined level is substantially equal to N−1.4. The method as defined in claim 1 further including the step oftransmitting to each mobile station the adaptive transmitter poweramplifier gain levels associated with the transmitter for that mobilestation.
 5. The method as defined in claim 4 further comprising the stepof adjusting the controllable gain amplifier gain levels at each mobilestation to a level defined by the associated adaptive transmitter poweramplifier gain levels in the transmitting step.
 6. The method as definedin claim 4 further comprising the step of supplying to the base stationreceivers the adaptive receiver power amplifier gain levels associatedwith the receiver of signals from that mobile station.
 7. The method asdefined in claim 6 further comprising the step of adjusting thecontrollable gain amplifier gain levels at each adaptive receiver in thebase station to a level defined by the associated adaptive receiverpower amplifier gain levels in the transmitting step.
 8. The method asdefined in claim 4 wherein the transmitting step further comprises thesteps of: associating the adaptive transmitter power amplifier gainlevels from the determining step with a particular mobile station; andassociating the adaptive receiver power amplifier gain levels from thedetermining step with a particular receiver in the base station. 9.Method for power adaptation in a multicarrier code division multipleaccess communication system in which a plurality of mobile stationscommunicate with at least one base station, each mobile stationincluding an adaptive transmitter that includes a plurality ofcontrollable gain amplifiers, the at least one base station including aplurality of adaptive receivers, each receiver corresponding to one ofthe plurality of mobile stations and including a plurality ofcontrollable gain amplifiers, the method comprising the steps of:receiving signals from the mobile stations; estimating channel powergain for each mobile station in response to the received mobile stationsignals, where the channel power gain G_(k,n) is for an n^(th)subcarrier of mobile station k, for k=1, 2, . . . , K and n=1, 2, . . ., N; ordering received signals from the mobile stations on a decreasingorder basis for the corresponding estimated channel power gain from themobile station associated with a highest channel gain to the mobilestation associated with a lowest channel gain; determining both aplurality of power amplifier gain levels α_(k,n) ^(t) for thecontrollable gain amplifiers in the adaptive transmitter associated witheach mobile station and a plurality of power amplifier gain levelsβ_(k,n) ^(t) for the controllable gain amplifiers in the adaptivereceiver associated with each mobile station in response to the orderedchannel power gains, wherein the adaptive transmitter power amplifiergain levels and the adaptive receiver power amplifier gain levels aredynamically adapted to the estimated channel power gains for each symbolfrom the mobile stations in the frequency domain for distributingavailable transmission power over a subset N′ of all N subcarriers,wherein 1≦N′≦N and the subset of subcarriers exhibits the N′ highestestimated channel power gains.
 10. The method as defined in claim 9wherein the determining step further includes the steps of: calculatingthe adaptive transmitter power amplifier gain levels as follows,$\alpha_{k,n}^{f} = \{ {\begin{matrix}{{1/N^{\prime}},} & {{{if}\mspace{14mu} G_{k,n}} > G_{k,N^{\prime}}^{\prime}} \\{0,} & {otherwise}\end{matrix},} $ where G_(k,N′) ^(t) an order statistic obtainedfrom the ordering step; and calculating the adaptive receiver poweramplifier gain levels as follows:$\beta_{k,n}^{f} = \{ {\begin{matrix}{G_{k,n},} & {{{if}\mspace{14mu} G_{k,n}} > G_{k,N^{\prime}}^{\prime}} \\{0,} & {otherwise}\end{matrix}.} $
 11. The method as defined in claim 9 furtherincluding the step of transmitting to each mobile station the adaptivetransmitter power amplifier gain levels associated with the transmitterfor that mobile station.
 12. The method as defined in claim 11 furthercomprising the step of adjusting the controllable gain amplifier gainlevels at each mobile station to a level defined by the associatedadaptive transmitter power amplifier gain levels in the transmittingstep.
 13. The method as defined in claim 11 further comprising the stepof supplying to the base station receivers the adaptive receiver poweramplifier gain levels associated with the receiver of signals from thatmobile station.
 14. The method as defined in claim 13 further comprisingthe step of adjusting the controllable gain amplifier gain levels ateach adaptive receiver in the base station to a level defined by theassociated adaptive receiver power amplifier gain levels in thetransmitting step.
 15. The method as defined in claim 11 wherein thetransmitting step further comprises the steps of: associating theadaptive transmitter power amplifier gain levels from the determiningstep with a particular mobile station; and associating the adaptivereceiver power amplifier gain levels from the determining step with aparticular receiver in the base station.
 16. Method for power adaptationin a multicarrier code division multiple access communication system inwhich a plurality of mobile stations communicate with at least one basestation, each mobile station including an adaptive transmitter thatincludes a plurality of controllable gain amplifiers, the at least onebase station including a plurality of adaptive receivers, each receivercorresponding to one of the plurality of mobile stations and including aplurality of controllable gain amplifiers, the method comprising thesteps of: receiving signals from the mobile stations; estimating channelpower gain for each mobile station in response to the received mobilestation signals, where the channel power gain G_(k,n) is for an n^(th)subcarrier of mobile station k, for k=1, 2, . . . , K and n=1, 2, . . ., N; ordering received signals from the mobile stations on a decreasingorder basis for the corresponding estimated channel power gain from themobile station associated with a highest channel gain to the mobilestation associated with a lowest channel gain; determining both aplurality of power amplifier gain levels α_(k,n) ^(t) for thecontrollable gain amplifiers in the adaptive transmitter associated witheach mobile station and a plurality of power amplifier gain levelsβ_(k,n) ^(t) for the controllable gain amplifiers in the adaptivereceiver associated with each mobile station in response to the orderedchannel power gains, wherein the adaptive transmitter power amplifiergain levels and the adaptive receiver power amplifier gain levels aredynamically adapted to the estimated channel power gains for each symbolfrom the mobile stations in the frequency domain for distributingavailable transmission power over a subset N′ of all N subcarriers,wherein 1≦N′≦N and the subset of subcarriers exhibits the N′ highestestimated channel power gains; and then revising the determination ofboth the determined plurality of power amplifier gain levels α_(k,n)^(t) for the controllable gain amplifiers in the adaptive transmitterassociated with each mobile station and the determined plurality ofpower amplifier gain levels β_(k,n) ^(t) for the controllable gainamplifiers in the adaptive receiver associated with each mobile stationin response to the ordered channel power gains, wherein the adaptivetransmitter power amplifier gain levels and the adaptive receiver poweramplifier gain levels are dynamically adapted to the estimated channelpower gains for each symbol from the mobile stations in the time domainfor distributing available transmission power over the subset of N′subcarriers to maintain a received signal strength for signals receivedat the base station at a predetermined level.
 17. The method as definedin claim 16 wherein the determining step further includes the steps of:calculating the adaptive transmitter power amplifier gain levels asfollows,α_(k,n) ^(t)=α_(k) ^(t) /N, n=1,2, . . . , N, where${\alpha_{k}^{t} = {G_{R}^{t}/G_{k}}},{G_{k} = {\sum\limits_{n = 0}^{N - 1}\; G_{k,n}}}$and G_(R) ^(t) is the predetermined level; and calculating the adaptivereceiver power amplifier gain levels as follows:β_(k,n) ^(t) =G _(k,n) , n=1,2, . . . , N.
 18. The method as defined inclaim 17 wherein the revising step further includes the steps of:calculating the adaptive transmitter power amplifier gain levels asfollows, $\alpha_{k,n}^{f} = \{ {\begin{matrix}{{1/N^{\prime}},{{{if}\mspace{14mu} G_{k,n}} > G_{k,N^{\prime}}^{\prime}}} \\{0,{otherwise}}\end{matrix},} $ where G_(k,N′) ^(t) an order statistic obtainedfrom the ordering step; and calculating the adaptive receiver poweramplifier gain levels as follows:$\beta_{k,n}^{f} = \{ {\begin{matrix}{G_{k,n},{{{if}\mspace{14mu} G_{k,n}} > G_{k,N^{\prime}}^{\prime}}} \\{0,{otherwise}}\end{matrix}.} $
 19. The method as defined in claim 18 whereinthe predetermined level is substantially equal to N−1.
 20. The method asdefined in claim 16 further including the step of transmitting to eachmobile station the adaptive transmitter power amplifier gain levelsassociated with the transmitter for that mobile station.
 21. The methodas defined in claim 20 further comprising the step of adjusting thecontrollable gain amplifier gain levels at each mobile station to alevel defined by the associated adaptive transmitter power amplifiergain levels in the transmitting step.
 22. The method as defined in claim20 further comprising the step of supplying to the base stationreceivers the adaptive receiver power amplifier gain levels associatedwith the receiver of signals from that mobile station.
 23. The method asdefined in claim 22 further comprising the step of adjusting thecontrollable gain amplifier gain levels at each adaptive receiver in thebase station to a level defined by the associated adaptive receiverpower amplifier gain levels in the transmitting step.
 24. The method asdefined in claim 20 wherein the transmitting step further comprises thesteps of: associating the adaptive transmitter power amplifier gainlevels from the determining step with a particular mobile station; andassociating the adaptive receiver power amplifier gain levels from thedetermining step with a particular receiver in the base station.
 25. Amethod for power adaptation in a multicarrier code division multipleaccess communication system, the method comprising: estimating channelpower gains for a multiplicity of sub-carriers for a multiplicity ofsignals received from a multiplicity of transmitting stations, to obtainestimated channel power gains; calculating one or more sets oftransmitter power gains for transmitter amplifiers of each sub-carrierassociated with one or more of the transmitting stations, using theestimated channel power gains, wherein said calculating includes atleast one type of calculating technique selected from at least one of: atime-domain technique comprising distributing transmitter power over allsub-carriers available to a particular transmitting station and adaptingtransmitter power for each transmitted symbol on each sub-carrier tomaintain a desired receiver signal strength; or a frequency-domaintechnique comprising distributing transmitter power over a subset of thesub-carriers available to a particular transmitting station, the subsetcomprising a set of sub-carriers available to the particulartransmitting station and having highest estimated channel power gains;and feeding back the one or more calculated sets of transmitter powergains to the respective transmitting stations.
 26. The method accordingto claim 25, wherein said calculating further comprises: calculating atleast one set of receiver amplifier gains based on the estimated channelpower gains, in conjunction with calculating said transmitter powergains.
 27. The method according to claim 25, wherein saidfrequency-domain technique further comprises: ordering the estimatedchannel power gains.
 28. The method according to claim 25, wherein saidfrequency-domain technique comprises: computing the transmitter powergains for the sub-carriers of a particular transmitting station suchthat all of the transmitter power gains are equal for sub-carriers inthe subset and are zero for sub-carriers not in the subset.
 29. Themethod according to claim 28, wherein said frequency-domain techniquefurther comprises: calculating at least one set of receiver amplifiergains based on the estimated channel power gains, in conjunction withcalculating said transmitter power gains, wherein the receiver amplifiergains are equal for sub-carriers in the subset and are zero forsub-carriers not in the subset.
 30. The method according to claim 25,wherein said time-domain technique further comprises: for a particulartransmitting station, and for each associated sub-carrier, computing atransmitter amplifier gain for each symbol based at least one a ratio ofsaid desired receiver signal strength to a sum of the estimated channelpower gains of the sub-carriers associated with the particulartransmitting station.
 31. The method according to claim 30, wherein saidcomputing a transmitter amplifier gain comprises: setting all of thetransmitter amplifier gains for the particular transmitting station tobe equal to each other for each symbol transmitted by the particulartransmitting station.
 32. An apparatus, comprising: a channel estimationmodule to estimate channel power gains for a multiplicity ofsub-carriers for a multiplicity of signals received from a multiplicityof transmitting stations, to obtain estimated channel power gains; again calculation module to calculate one or more sets of transmitterpower gains for transmitter amplifiers of each sub-carrier associatedwith one or more of the transmitting stations, using the estimatedchannel power gains, wherein the gain calculation module uses at leastone type of calculating technique selected from at least one of: atime-domain technique comprising distributing transmitter power over allsub-carriers available to a particular transmitting station and adaptingtransmitter power for each transmitted symbol on each sub-carrier tomaintain a desired receiver signal strength; or a frequency-domaintechnique comprising distributing transmitter power over a subset of thesub-carriers available to a particular transmitting station, the subsetcomprising a set of sub-carriers available to the particulartransmitting station and having highest estimated channel power gains;and a feedback channel to feed back the one or more calculated sets oftransmitter power gains to the respective transmitting stations.
 33. Theapparatus according to claim 32, wherein said gain calculation module isfurther to calculate at least one set of receiver amplifier gains basedon the estimated channel power gains, in conjunction with calculatingsaid transmitter power gains.
 34. The apparatus according to claim 32,wherein said frequency-domain technique further comprises: ordering theestimated channel power gains.
 35. The apparatus according to claim 32,wherein said frequency-domain technique comprises: computing thetransmitter power gains for the sub-carriers of a particulartransmitting station such that all of the transmitter power gains areequal for sub-carriers in the subset and are zero for sub-carriers notin the subset.
 36. The apparatus according to claim 35, wherein saidfrequency-domain technique further comprises: calculating at least oneset of receiver amplifier gains based on the estimated channel powergains, in conjunction with calculating said transmitter power gains,wherein the receiver amplifier gains are equal for sub-carriers in thesubset and are zero for sub-carriers not in the subset.
 37. Theapparatus according to claim 32, wherein said time-domain techniquefurther comprises: for a particular transmitting station, and for eachassociated sub-carrier, computing a transmitter amplifier gain for eachsymbol based at least one a ratio of said desired receiver signalstrength to a sum of the estimated channel power gains of thesub-carriers associated with the particular transmitting station. 38.The apparatus according to claim 37, wherein said computing atransmitter amplifier gain comprises: setting all of the transmitteramplifier gains for the particular transmitting station to be equal toeach other for each symbol transmitted by the particular transmittingstation.